Wideband Tunable Frequency Single-Sideband Converter with PVT Tracking

ABSTRACT

A wideband tunable frequency single-subband converter is proposed. The wideband frequency tunable converter operates within a wideband and tunable frequency range, and has process, voltage, and temperature (PVT) tracking capability. In one embodiment, the wideband converter comprises a frequency tunable polyphase filter having a plurality of switchable polyphase resistors. The polyphase resistors are controlled by a frequency tuning control signal to achieve wideband frequency tunability. In a preferred embodiment, a triode mode transistor is used as a polyphase resistor, and a different resistance value of the polyphase filter is realized by turning on one or multiple of the different transistors in triode mode. In addition, a constant Gm(R) bias generator is used to provide the gate biases to the triode mode transistors to maintain a constant and stable resistance value across PVT and other variation.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims priority under 35 U.S.C. § 119 from U.S.Provisional Application No. 62/940,716, entitled “Wideband TunableFrequency Single-Sideband Converter with PVT Tracking,” filed on Nov.26, 2019, the subject matter of which is incorporated herein byreference.

TECHNICAL FIELD

The disclosed embodiments relate generally to wireless networkcommunications, and, more particularly, to frequency tunable converterwith process, voltage, and temperature (PVT) tracking.

BACKGROUND

Radio Frequency (RF) converters are integrated component assembliesrequired for converting microwave signals into lower (or intermediate)or higher frequency ranges for further processing. They generallyconsist of an input filter, a local oscillator filter, an IF filter, amixer, and frequently an LO frequency multiplier, plus one or morestages of IF amplification. RF frequency converter; may also incorporatea local oscilator, gain compensation (GC) components, and an RFpreamplifier. As a system, RF frequency converters function to alterincoming microwave signals into different frequency ranges to allow fora wide range of processing options that could follow. RF frequencyconverters are available in a number of configurations, defined by thetype of frequency they output. Upconverters change microwave signals toa. higher frequency range. Generally an upconverter is designed toproduce an output signal frequency for a particular frequency band. Bycontrast, downconverters alter microwave. signals in to an. intermediatefrequency (IF) range, again tuned to a particular frequency band. Somevarieties of RF frequency converters are dual upconverters anddownconverters, meaning that they can modulate the frequency either upor down, but again, only into a specific range on either side of thespectrum. A final type of converter is the variable converter, which canchange the frequency of the input signal to any frequency within theoperating range. They are not constrained to produce signals for aparticular frequency band, as is the case with upconverters anddownconverters.

Wideband tunable frequency converters can operate on wideband, whiletuning its operating frequency. RF converters configured using radiofrequency integrated circuit (RFIC) are subject to 1) PVT variation(variations in the wafer process, supply voltage, andtemperature)—typically results in several dBs of variations ifuncompensated; and 2) random variations due to transistor or passiveelement size variations—this requirement is usually met by limiting thesmallest size of transistor, capacitor, resistor to be used within theRFIC. Solutions are sought to achieve wideband frequency tunability, andwith PVT tracking to improve the performance of wideband tunablefrequency converters.

SUMMARY

A wideband tunable frequency single-subband converter is proposed. Thewideband frequency tunable converter operates within a wideband andtunable frequency range, and has process, voltage, and temperature (PVT)tracking capability. In one embodiment, the wideband converter comprisesa frequency tunable polyphase filter having a plurality of switchablepolyphase resistors. The polyphase resistors are controlled by afrequency tuning control signal to achieve wideband frequencytunability. In a preferred embodiment, a triode mode transistor is usedas a polyphase resistor, and a different resistance value of thepolyphase filter is realized by turning on one or multiple of thedifferent transistors in triode mode. In addition, a constant Gm(R) biasgenerator is used to provide the gate biases to the triode modetransistors to maintain a constant and stable resistance value acrossPVT and other variation.

In one embodiment, a wideband frequency tunable converter with PVTtracking receives an input signal (IF) having a frequency of f_(IF) by awideband frequency tunable polyphase filter. The polyphase filterconverts the IF signal to IF_(I) and IF_(Q). The converter amplifiesIF_(I) and IF_(Q) by a pair of wideband frequency tunable amplifiers andthereby generating amplified input signals with or without polarityinversion of IF_(I) and IF_(Q). The converter multiplies the amplifiedinput signals with local oscillator (LO) signals by a pair of doublesideband mixers, the LO signals having a frequency of f_(LO). Theconverter outputs an output signal (RF) from an output summer that arecoupled to the mixers. The RF signal has an image frequency of(f_(IF)+f_(LO)) or (f_(IF)−f_(LO)) under up conversion, selectable bythe polarity inversion.

In one preferred embodiment, the polyphase filter is a complex domainfilter which comprises a single or multiple stages, each stage containsa set of cross connected resistor and capacitor pairs that converts asingle-ended or a differential signal into quadrature differentialsignals at a specific frequency or vice versa. A wideband polyphasefilter can be realized with switchable of resistor of different values,controlled by a frequency tuning control signal. To overcome theresistance variations due to PVT, the polyphase resistor comprises aplurality of triode mode transistors having different transistor sizes,and a gate voltage of each triode mode transistor is provided by aconstant Gm(R) bias generator. The constant Gm(R) bias generatorprovides the bias such that the resistance value of each triode modetransistor remains stable over process, voltage, and temperature (PVT)variation.

Other embodiments and advantages are described in the detaileddescription below. This summary does not purport to define theinvention. The invention is defined by the claims.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified circuit diagram of a wideband tunable frequencysingle-sideband converter in accordance with one novel aspect.

FIG. 2 is a simplified circuit diagram of a single-sidebandsingle-balanced converter that illustrates the operation of asingle-sideband mixer.

FIG. 3 illustrates one embodiment of a configurable signal polarityinverter and wideband frequency tunable amplifier.

FIG. 4 illustrates a simplified circuit diagram of a wideband frequencytunable polyphase filter in accordance with one novel aspect.

FIG. 5 illustrates a preferred embodiment of a switched-R inside apolyphase filter.

FIG. 6 illustrates a preferred embodiment of a switch for controllingpolyphase resistors inside a polyphase filter.

FIG. 7 illustrates a preferred embodiment of a constant Gm(R) biasgenerator for providing constant bias voltage to switches that controlpolyphase resistors inside a polyphase filter.

FIG. 8 illustrates the image level over phase or amplitude error in asingle-sideband mixer conversion.

FIG. 9A illustrates RC time constant calibration used for polyphasefilter.

FIG. 9B illustrates one embodiment of RC time constant calibrationprocedure on state increment of switch-R bank.

FIG. 10 is a flow chart of a method of converting wideband radiofrequency signals with tunable frequency and PVT tracking by a widebandconverter in accordance with one novel aspect.

DETAILED DESCRIPTION

Reference will now be made in detail to some embodiments of theinvention, examples of which are illustrated in the accompanyingdrawings.

FIG. 1 is a simplified circuit diagram of a wideband tunable frequencysingle-sideband converter 100 in accordance with one novel aspect.Wideband tunable frequency single-sideband converter 100 comprises awideband frequency tunable polyphase filter 110, two configurable signalpolarity inverter 120, two wideband frequency tunable amplifier 130, twodouble sideband mixers 140, and an output summer 150. In the example ofFIG. 1, the input signal IF has a frequency of f_(IF) having a widebandfrequency range of 50-7000 Hz, the local oscillator signal LO has afrequency of f_(LO). The output signal RF has an image frequency that iseither (f_(IF)+f_(LO)) or (f_(IF)−f_(LO)) for up conversion, dependingon whether the polarity inverter 120 is polarity non-inversion (lowersideband mode) or polarity inversion (higher sideband mode).

In accordance with one novel aspect, the wideband frequency tunableconverter 100 operates within a wideband and tunable frequency range,and has process, voltage, and temperature (PVT) tracking capability. Inone embodiment, the wideband frequency tunable polyphase filter 110comprises a plurality of switchable polyphase resistors. The polyphaseresistors are controlled by a frequency tuning control signal to achievewideband frequency tunability, e.g., a resistor switch control signal160 in FIG. 1. In a preferred embodiment, a triode mode transistor isused as a polyphase resistor, and a different resistance value of thepolyphase filter is realized by turning on one or multiple of thedifferent transistors in triode mode. In addition, a constant Gm(R) biasgenerator is used to provide the gate biases to the triode modetransistors to maintain a constant and stable resistance value acrossPVT variation.

The operation principle of the single-sideband double-balanced convertercan be explained using a simplified single-sideband single-balancedconverter. FIG. 2 is a simplified circuit diagram of a single-sidebandsingle-balanced converter 200 that illustrates the operation of asingle-sideband mixer. Single-sideband single-balanced converter 200comprises two frequency tunable amplifiers 230, a single-balanced mixer240, and a output summer 250. In the example of FIG. 2, the input signalIF has a frequency of f_(IF), the local oscillator signal LO has afrequency of f_(LO). The output signal RF has two image frequencies thatis either (f_(IF)+f_(LO)) and (f_(LO)−f_(IF)) for up conversion,selectable by the signal polarity inversion. For down conversion, thesignal enters from opposite direction (marked as RF) in FIG. 2 and theimage IF frequency is (f_(RF)−f_(LO))or (f_(LO)−f_(RF)), selectable bythe signal polarity inversion.

The differential signal input IF_(I) and IF_(Q) are quadraturedifferential signals differ in phase by 90 degree:

IF _(I)=(COS(2πf _(IF) t), −COS(2πf _(IF) t))

IF _(Q)=(SIN(2πf _(IF) t), −SIN (2πf _(IF) t))

The configurable signal polarity inverter in frequency tunableamplifiers 230 is to invert the signal polarity or leave the signalpolarity unchanged of IF_(I) and IF_(Q):

+/−IF _(I)=+/−1*(COS(2πf _(IF) t), −COS(2πf _(IF) t))

+/−IF _(Q)=+/−1*(SIN(2πf _(IF) t), −SIN(2πf _(IF) t))

The LO signals come from a local oscillator and contain quadraturedifferential signals LO_(I) and LO_(Q) that differ in phase by 90degree:

LO _(I)=(COS(2πf _(LO) t), −COS(2πf _(LO) t))

LO _(Q)=(SIN(2πf _(LO) t), −SIN(2πf _(LO) t))

To remove one of the image frequency out of the upconverter, bothin-phase and quadrature phase components are needed for both input andLO frequencies. The single-balanced mixer 240 and output summer 250operation can be represented by equation (1) when polarity inverter in230 is polarity non-inversion (lower sideband mode), and equation (2)when polarity inverter in 230 is polarity inversion (higher sidebandmode):

$\begin{matrix}\begin{matrix}{{RF} = \begin{pmatrix}{{{{{COS}\left( {2\pi \; f_{IF}t} \right)}*{{COS}\left( {2\pi \; f_{LO}t} \right)}} + {{{SIN}\left( {2\pi \; f_{IF}t} \right)}*{SIN}\left( {2\pi \; f_{LO}t} \right)}},} \\{{{- {{COS}\left( {2\pi \; f_{IF}t} \right)}}*{{COS}\left( {2\pi \; f_{LO}t} \right)}} - {{{SIN}\left( {2\pi \; f_{IF}t} \right)}*{{SIN}\left( {2\pi \; f_{LO}t} \right)}}}\end{pmatrix}} \\{= \left( {{{COS}\left( {2{\pi \left\lbrack {f_{IF} - f_{LO}} \right\rbrack}t} \right)},{- {{COS}\left( {2{\pi \left\lbrack {f_{IF} - f_{LO}} \right\rbrack}t} \right)}}} \right)}\end{matrix} & (1) \\\begin{matrix}{{RF} = \begin{pmatrix}{{{{{COS}\left( {2\pi \; f_{IF}t} \right)}*{{COS}\left( {2\pi \; f_{LO}t} \right)}} - {{{SIN}\left( {2\pi \; f_{IF}t} \right)}*{SIN}\left( {2\pi \; f_{LO}t} \right)}},} \\{{{- {{COS}\left( {2\pi \; f_{IF}t} \right)}}*{{COS}\left( {2\pi \; f_{LO}t} \right)}} + {{{SIN}\left( {2\pi \; f_{IF}t} \right)}*{{SIN}\left( {2\pi \; f_{LO}t} \right)}}}\end{pmatrix}} \\{= \left( {{{COS}\left( {2{\pi \left\lbrack {f_{IF} + f_{LO}} \right\rbrack}t} \right)},{- {{COS}\left( {2{\pi \left\lbrack {f_{IF} + f_{LO}} \right\rbrack}t} \right)}}} \right)}\end{matrix} & (2)\end{matrix}$

FIG. 3 illustrates one embodiment of a configurable signal polarityinverter and wideband frequency tunable amplifier 300. The amplifier 300is an exemplary embodiment of the configurable signal polarity inverter120 and the wideband frequency tunable amplifier 130 as depicted in FIG.1, which can be used in the wideband tunable frequency single-sidebandconverter 100 as illustrated in FIG. 1. In the embodiment of FIG. 3, theinput radio frequency signal RFin is coupled to an input transformer310, and the output radio frequency signal RFout is coupled to an outputtransformer 320. Stacked transformers 310 and 320 are controller by bandswitches 311 and 312 (switched C), respectively, to achieve thefrequency tunability of amplifier 300. Different values of the switchedC and the transform resonates achieving a match condition at input andoutput at different frequencies. Amplifier 300 comprises a maintransconductance pair 330, which is formed by two complementarydifferential transistor pairs. To achieve phase inverter of theamplifier, a phase switch 340 is used to select which of thecomplementary differential pairs is turned on or turned off. As aresult, the polarity of the input signal RFin can be inverted or remainthe same. Furthermore, self-neutralization of the drain-to-gatecapacitance, e.g., Cgd, in transistors provides excellent reverseisolation and stability. Without neutralization of the parasiticcapacitance Cgd, the RF signal would leak to the output from the inputor leak to the input from the output. With Self-neutralization, iteliminates the drain-to-gate capacitance via negative feedbacks, i.e.,controlled amount of cross-connection to the opposite input signalpolarity.

FIG. 4 illustrates a simplified circuit diagram of a wideband frequencytunable polyphase filter 400 in accordance with one novel aspect.Polyphase filter 400 is an exemplary embodiment of the widebandfrequency tunable polyphase filter 110 as depicted in FIG. 1, which canbe used in the wideband tunable frequency single-sideband converter 100as illustrated in FIG. 1. In the embodiment of FIG. 4, polyphase filter400 generates four quadrant (polyphase) signals, e.g., (I, Q, I_bar,Q_bar), or four signals with (0°, 90°, 180°, 270°) phase and equalmagnitude. Note that the I and Q signals are (0°, 90°), and the othertwo signals I_bar and Q_bar are (180°, 270°). The four output signals(0°, 90°, 180°, 270°) are quadrature signals.

The polyphase filter 400 comprises a plurality of R-C networks,consisting of a plurality of polyphase resistors and capacitors that canproduce polyphase signals. The R and C time constant determines theoperating frequency. To achieve wideband frequency tunability, eachpolyphase resistor is switchable, as controlled by a frequency tuningcontrol signal. Each polyphase resistor is referred to as a switched-R,as depicted by 410 conceptually. The switched-R 410 comprises fourparallel resistors R1, R2, R3, and R4, each controlled by a switch C1,C2, C3, and C4, respectively. By controlling the different switches,different resistor values of polyphase filter 400 can be realized. Inone example, a four-bit frequency tuning control signal, each bitcontrolling one of the four switches, can control up to 2⁴=16possibilities of the corresponding resistance value of the polyphaseresistor 410.

FIG. 5 illustrates a preferred embodiment of a polyphase resistor(switched-R) 510 inside a polyphase filter. In the embodiment of FIG. 5,switched-R 510 comprises four triode mode MOSFET transistors, suppliedby a constant Gm(R) bias generator 520. A MOSFET is said to operate inthree regions, cutoff, triode and saturation, based on the condition ofthe inversion layer existed between the source and drain, as depicted inI-V curve 530. The triode region is the operating region where theinversion region exists and current flows, but this region has begun totaper near the source. The potential requirement here is Vds<Vgs−Vth.Here, the drain source current has a parabolic relationship with thedrain source potential. The MOSFET can simultaneously operate as aswitch, in the “off” mode when it is turned off and in the “on” modewhen it is at the triode region. The linear region of a MOSFET can beconsidered as a special portion of the triode region, where because ofthe very small value of the applied drain-source potential, there is aroughly linear relationship between Vds and Ids and the MOSFET behaveslike a voltage dependent resistor. The potential condition for thelinear region or the “deep triode” region is Vds<<Vgs−Vth.

As depicted in FIG. 5, different resistor value R of the switched-R 510is realized by turning on one or more of the different transistors intriode mode, each transistor with a different Gm value (depending on thecorresponding transistor size). That is, R₁=(1/Gm1), R₂=(1/Gm2),R₃=(1/Gm3), and R₄=(1/Gm4). For example, if C1 and C2 are on, thenR=R₁∥R₂, if C2 and C3 are on, then R=R₂∥R₃. In order to maintain each R(or Gm) value across process, voltage, and temperature (PVT) variations,a constant Gm(R) bias voltage generator 520 is used to provide the gatebiases V_(GS)(C1, C2, C3, C4) to the triode mode transistors.

FIG. 6 illustrates a preferred embodiment of a switch 610 forcontrolling polyphase resistors inside a polyphase filter. Asillustrated earlier in FIG. 4, a polyphase resistor (e.g., 410) is aswitched-R, which comprises four parallel resistors R1, R2, R3, and R4,each resistor is controlled by a switch C1, C2, C3, and C4,respectively. In the embodiment of FIG. 6, switch 610 (C4) is controlledby a constant Gm(R) bias generator 620, which generate a constant biasvoltage V_(GS). The MOSFET can simultaneously operate as a switch, it isin the “off” mode when it is turned off and in the “on” mode when it isin triode region.

In a typical semiconductor process, the resistor value variessignificantly with process and temperature variations. FIG. 7illustrates a preferred embodiment of a constant Gm(R) bias generator710 for providing constant bias voltage V_(GS) to switches that controlpolyphase resistors inside a polyphase filter. The structure of theconstant Gm(R) bias generator circuit 710 has three parts: the replicatransistor 711, the operation amplifier (OPA) 712, and the bandgapcircuit 713. The replica transistor size could be the same or scaledwith the transistor size of the switched-R. The current referenceI_(Reference) of the bandgap circuit 713 defines the drain to sourcecurrent I_(DS) of the replica transistor 711. The negative feedbackconnected OPA 712 forces the drain voltage V_(DS) of the replicatransistor 711 to be equal to the voltage reference V_(Reference) of thebandgap circuit 713 by lifting or lowing the gate voltage V_(GS) of thereplica transistor 711. The equilibrium biasing of the replicatransistor 711 will be V_(DS)=V_(reference) and I_(DS)=I_(Reference).Since the ratio of voltage and the current from the same bandgap circuit713 are all very stable over process, voltage, and temperature (PVT)variation, the channel resistance (V_(DS)/I_(DS)) of the replicatransistor 711 is a constant. The V_(GS) is then applying to otherswitched-R transistors in a polyphase filter that would make them alsobe a constant channel resistance over process, voltage, and temperature(PVT) and other corner variation. To design the value of the channelresistance in a polyphase filter, a transistor size that is n times overthe replica transistor size can be chosen. For example, the channelresistance of the chosen transistor is (V_(DS)/I_(DS))/n. If n=2, thenthe channel resistance in polyphase filter is (V_(DS)/I_(DS))/2.

FIG. 8 illustrates the image level over phase or amplitude error in asingle-sideband mixer conversion, as depicted by 810. The phase errorand amplitude error would cause the system impairment that the imagewould occur in single-side-band mixer conversion. The image leveldepends on how good is the phase error and amplitude error is, as shownin FIG. 8. The polyphase filter will have perfect phase error andamplitude error at corner frequency Fc. To achieve frequency tunablepolyphase filter, the corner frequency Fc, either R or C in thepolyphase filter is adjusted. In a preferred embodiment, as illustratedearlier, the R is adjusted via triode transistor. The R is regulated bya constant Gm(R) bias generator.

FIG. 9A illustrates RC time constant calibration used for polyphasefilter. For R-C circuits, the R is regulated by const-Gm(R) biasgenerate over process, voltage and temperature (PVT) variation. However,despite the capacitor C is insensitive to temperature, the capacitor Cstill has process variation. The RC time constant calibration is tocalibrate the Fc of polyphase filter over the process variation and onlyhave to do it one time after the chip is made. In FIG. 9A, the preciseclock 902 generates two period T, amplitude Vpeak, but opposite phasesquare waves to feed in the switched-capacitor resistor. The equivalentresistance of switched-capacitor resistor 903 is T/C. The R banks,switched-capacitor resistor is voltage divider for precise DC reference.R1, R2 is another voltage divider for precise DC reference. The bypasscapacitor is to attenuate the clock feedthrough at node X.

The level of precise DC reference should make sure the triode transistoris always valid in the switch-R bank. In other words, Vref−V_(x)<Vov(overdrive voltage of the transistor). The value R and C can be scaledup or down comparing to the value R and C of the polyphase filter. Theperiod T/2 should be several RC time constant and it depends on howaccuracy the calibration is needed. For example if T/2 large than 3 RCtime constant, then the error can be smaller than 5%.

Error(%)=100×(1−e ^(−T/2RC))

The calibration algorithm is as follows. The goal is to search theclosest RC/T value to R1/R2 value. The R can be searched from smallestto largest on each certain several cycle of the precise clock, or eitherway. Once the DC comparator 905 flip its output sign from low to high,the precise clock will stop and the calibration is completed. Thepolarity of DC comparator, in other words, V_(X)-V_(Y) or V_(Y)-V_(X),depends on the direction of searching R in R banks. For example, if theR is searched from smallest to largest, the polarity of DC comparatorwill be V_(X)-V_(Y), vice versa.

${{Search}\mspace{14mu}\left\lbrack \frac{R_{i}C}{T} \right\rbrack}{_{{i = {0\mspace{14mu} {to}\mspace{14mu} n}},{{{or}\mspace{14mu} i} = {n\mspace{14mu} {to}\mspace{14mu} 0}}}\mspace{14mu} {{{closest}\mspace{14mu} {to}\mspace{14mu} \frac{R_{1}}{R_{2}}},}}$

Where

-   n is the array number in R banks.

FIG. 9B illustrates one embodiment of RC time constant calibrationprocedure on state increment of switch-R bank. In the example of FIG.9B, a 4-bit state increment embodiment is shown by 910 and 920. Theprecise clock 902 will generate a T2 period clock and will be large thanT. The longer the T2 period is, the slower the DC comparator can beused. The slower DC comparator means the more accuracy can be achieved.The 4-bit counter 911 will change to the next state when eachfalling-edge arrive at the counter. If R banks has n-bits triodetransistor, then the counter will be n-bits.

FIG. 10 is a flow chart of a method of converting wideband radiofrequency signals with tunable frequency and PVT tracking by a widebandconverter in accordance with one novel aspect. In step 1001, theconverter receives an input signal (IF) having a frequency of f_(IF) bya wideband frequency tunable polyphase filter. The polyphase filterconverts the IF signal to IF_(I) and IF_(Q). In step 1002, the widebandconverter amplifies IF_(I) and IF_(Q) by a pair of wideband frequencytunable amplifiers with signal polarity inverter and thereby generatingamplified input signals with or without a polarity inversion of IF_(I)and IF_(Q). In step 1003, the wideband converter multiplies theamplified input signals with local oscillator (LO) signals by a pair ofdouble sideband mixers, the LO signals having a frequency of f_(LO). Instep 1004, the wideband converter outputs an output signal (RF) from anoutput summer that are coupled to the mixers. The RF signal has an imagefrequency of (f_(IF)+f_(LO)) or (f_(IF)−f_(LO)) under up conversion,selectable by the polarity inversion.

Although the present invention has been described in connection withcertain specific embodiments for instructional purposes, the presentinvention is not limited thereto. Accordingly, various modifications,adaptations, and combinations of various features of the describedembodiments can be practiced without departing from the scope of theinvention as set forth in the claims.

What is claimed is:
 1. A wideband frequency tunable converter,comprising: a wideband frequency tunable polyphase filter that receivesan input signal (IF) having a frequency of f_(IF), wherein the polyphasefilter converts the IF signal to IF_(I) and IF_(Q); a pair of widebandfrequency tunable amplifiers that receive IF_(I) and IF_(Q) andgenerates amplified input signals with or without a polarity inversionof IF_(I) and IF_(Q); a pair of double sideband mixers that multiply theamplified input signals with local oscillator (LO) signals having afrequency of f_(LO); and an output summer that are coupled to the mixersand outputs an output signal (RF), wherein the RF signal has an imagefrequency of either (f_(IF)+f_(LO)) or (f_(IF)−f_(LO)) under upconversion, selectable by the polarity inversion.
 2. The converter ofclaim 1, wherein the converter operates under a tunable frequency andwith process, voltage, and temperature (PVT) tracking and compensation.3. The converter of claim 1, wherein the polyphase filter comprises aplurality of polyphase resistors that are controlled by a frequencytuning control signal.
 4. The converter of claim 3, wherein eachpolyphase resistor comprises a plurality of triode mode transistorshaving different transistor sizes.
 5. The converter of claim 4, whereina gate voltage of each triode mode transistor is provided by a constantGm(R) bias generator.
 6. The converter of claim 5, wherein the constantGm(R) bias generator comprises a replica transistor having a constantchannel resistance value over process, voltage, and temperature (PVT)variation.
 7. The converter of claim 6, wherein a resistance value ofeach triode mode transistor is determined based on a correspondingtransistor size and the channel resistance value of the replicatransistor.
 8. The converter of claim 6, wherein a resistance value ofeach triode mode transistor remains constant over process, voltage, andtemperature (PVT) variation.
 9. The converter of claim 1, wherein theamplifier comprises an input transformer and an output transformer witha band switch to achieve frequency tunability.
 10. The converter ofclaim 1, wherein the amplifier comprises a complementary differentialtransistor pair to achieve phase inversion of IF_(I) and IF_(Q).
 11. Theconverter of claim 1, wherein the polyphase filter comprises one ormultiple sets of cross connected resistor and capacitor pairs, whereinan RC time constant of each resistor and capacitor pair is calibratedusing a precise clock generator.
 12. A method for converting a widebandradio frequency (RF) signal with process, voltage, and temperature (PVT)tracking, comprising: receiving an input signal (IF) having a frequencyof f_(IF) by a wideband frequency tunable polyphase filter, wherein thepolyphase filter converts the IF signal to IF_(I) and IF_(Q); amplifyingIF_(I) and IF_(Q) by a pair of wideband frequency tunable amplifiers andthereby generating amplified input signals with or without a polarityinversion of IF_(I) and IF_(Q); multiplying the amplified input signalswith local oscillator (LO) signals by a pair of double sideband mixers,the LO signals having a frequency of f_(Lo; and) outputting an outputsignal (RF) from an output summer that are coupled to the mixers,wherein the RF signal has a frequency of (f_(IF)+f_(LO)) or(f_(IF)−f_(LO)) under up conversion, selectable by the polarityinversion.
 13. The method of claim 12, wherein the converter operatesunder a tunable frequency and with process, voltage, and temperature(PVT) tracking and compensation.
 14. The method of claim 12, wherein thepolyphase filter comprises a plurality of polyphase resistors that arecontrolled by a frequency tuning control signal.
 15. The method of claim14, wherein each polyphase resistor comprises a plurality of triode modetransistors having different transistor sizes.
 16. The method of claim15, wherein a gate voltage of each triode mode transistor is provided bya constant Gm(R) bias generator.
 17. The method of claim 16, wherein theconstant Gm(R) bias generator comprises a replica transistor having aconstant channel resistance value over process, voltage, and temperature(PVT) variation.
 18. The method of claim 17, wherein a resistance valueof each triode mode transistor is determined based on a correspondingtransistor size and the channel resistance value of the replicatransistor.
 19. The method of claim 17, wherein a resistance value ofeach triode mode transistor remains constant over process, voltage, andtemperature (PVT) variation.
 20. The method of claim 12, wherein theamplifier comprises an input transformer and an output transformer witha band switch to achieve frequency tunability.
 21. The method of claim12, wherein the amplifier comprises a complementary differentialtransistor pair to achieve phase inversion of IF_(I) and IF_(Q).
 22. Themethod of claim 12, wherein the polyphase filter comprises one ormultiple sets of cross connected resistor and capacitor pairs, whereinan RC time constant of each resistor and capacitor pair is calibratedusing a precise clock generator.